Dc to dc converter

ABSTRACT

A DC to DC converter includes an inductor having a first terminal and a second terminal, the first terminal coupled to an input of the DC to DC converter and the second terminal being coupled to an output of the DC to DC converter. A first switch is coupled between the second terminal and a current sensor. The switch controls current flow through the inductor and generates an inductor current signal representative of the current flow through the sensor. A slope generator generates a slope compensation signal. A first mixer adds the slope compensation signal to the inductor current signal. A sample and hold circuit samples a portion of the slope compensation signal. A second mixer subtracts the sampled portion of the slope compensation signal from the output of the first mixer, wherein inductor charging is terminated in response to the output of the first mixer.

This patent application claims priority to U.S. provisional patentapplication 61/847,348, filed on Jul. 17, 2013, entitled REDUCING THEVARIATION OF INDUCTOR CURRENT LIMIT WITH DUTY CYCLE BY REMOVING SLOPECOMPENSATION IN A CURRENT MODE CONTROLLED DC-DC CONVERTER, naming AjayK. Hari et al. as inventors, which is hereby incorporated by referencefor all purposes.

BACKGROUND

Some embodiments of DC to DC converters charge an inductor for a period.The inductor is then discharged into a capacitor. Discharging theinductor into the capacitor changes the voltage across the capacitor.The capacitor voltage is the output voltage, although in someembodiments, the capacitor voltage is filtered to remove noise and thefiltered voltage is the output voltage. The charging and discharging ofthe inductor is similar to a pulse width modulation system whereby thelengths of the charging and/or discharging determines the outputvoltage.

Some of the DC-DC converters use current mode control, which controlsthe output voltage based on the current passing through the inductor.Current mode control is susceptible to sub-harmonic oscillation for dutycycles greater than fifty percent in peak mode and duty cycles less thanfifty percent in valley mode. Peak mode is where the current peaks arecontrolled and valley mode is where current valleys are controlled. Thesub-harmonic oscillations cause noise and inaccurate DC to DCconversions.

SUMMARY

An embodiment of a DC to DC includes an inductor having a first terminaland a second terminal, the first terminal is coupled to an input of theDC to DC converter and the second terminal is coupled to an output ofthe DC to DC converter. A first switch is coupled between the secondterminal of the inductor and a current sensor, the first switch has aclosed state when current flows through the first switch and an openstate when current does not flow through the first switch. The currentsensor is coupled between the first switch and a node, wherein thecurrent sensor is for generating an inductor current signalrepresentative of the current flow through the sensor. A slope generatoris for generating a slope compensation signal. A first mixer having anoutput adds the slope compensation signal to the inductor currentsignal. A sample and hold circuit samples a portion of the slopecompensation signal. A second mixer having an output subtracts thesampled portion of the slope compensation signal from the output of thefirst mixer. The first switch is opened in response to the output of thefirst mixer.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a prior art DC to DC converter.

FIG. 2 is a schematic diagram of an embodiment of a DC to DC converterthat overcomes the problems associated with the converter of FIG. 1.

FIG. 3A is a graph depicting an embodiment of the inductor currentthrough the inductor of FIG. 2.

FIG. 3B is a graph depicting an embodiment of the slope compensationgenerated by the slope generator of FIG. 2.

FIG. 3C is a graph depicting the summation of the inductor current ofFIG. 3A and the slope compensation of FIG. 3B.

FIG. 3D is a graph depicting the subtraction of the peak slopecompensation from the graph of FIG. 3C.

FIG. 4 is a flowchart describing the operation of the DC to DC converterof FIG. 2.

DETAILED DESCRIPTION

Reference is made to FIG. 1, which is an embodiment of a prior art DC toDC converter 100 operating in a buck mode. The converter 100 typicallyincludes other circuitry and devices that monitor the output voltage.The devices and methods described herein are related to current sensing,so output voltage monitoring is not described with reference to FIG. 1.The converter 100 has an input 102 where an input voltage V_(IN) isapplied and an output 104 where an output voltage V_(OUT) is output fromthe converter 100. Both the input voltage V_(IN) and the output voltageV_(OUT) are DC voltages.

The converter 100 includes a switch, which in the embodiment of FIG. 1is a field effect transistor (FET) Q1 that is coupled to the input 102.Source current i_(S)(t) flows from the input 102 and through the FET Q1depending on the gate voltage of the FET Q1. A diode D1 and an inductorL1 are coupled (e.g. connected) to the output of the FET Q1. Theinductor L1 outputs a load current I_(L)(t) to a capacitor C1 and aresistor R1. The voltage across the resistor R1 and the capacitor C1 isthe output voltage V_(OUT). The on-time and off-time of the FET Q1determines the charging of the inductor L1, which determines thecharging of and voltage across the capacitor C1. As described above, thevoltage across the capacitor C1 is the output voltage V_(OUT) of theconverter 100.

The FET Q1 is controlled by a controller 110, which in the embodiment ofFIG. 1 is a latch 110. The output Q of the latch 110 controls the stateof the FET Q1, wherein the FET Q1 is either open or closed depending onthe gate voltage of the FET Q1. An input S of the latch 110 is coupledto a clock 112. The output Q is normally equal to the input S, unlessthe input R is set. As described below, the input R sets when the sourcecurrent I_(S)(t) is greater than a predetermined value.

A current sensor 120 measures the source current I_(S)(t) into theinductor L1. The current sensor 120 outputs a voltage that is input to acomparator 122. The comparator 122 compares the voltage output by thecurrent sensor 120 to a reference voltage V_(IREF). The voltage V_(IREF)is representative of the maximum current that should charge the inductorL1. When the current I_(S)(t) surpasses the current corresponding to thevoltage V_(IREF), the comparator 122 outputs a voltage that resets thelatch 112 and opens the FET Q1, which stops the inductor L1 fromcharging.

The converter 100 described above directly limits the current I_(L)(t)in the inductor L1 when the current I_(L)(t) is at a peak or when itexceeds a value corresponding to the voltage V_(IREF). For example, thecurrent I_(L)(t) is similar to the current I_(S)(t) when the inductor L1is charging. The converter 100 can be configured to limit the currentI_(L)(t) in the inductor L1 by limiting the valley current I_(L)(t) orpeak current I_(L)(t) as described above. Based on the foregoing, theinductor valley or peak current I_(L)(t) is limited directly, however,the average inductor current that is indirectly limited varies with theduty cycle of the current I_(L)(t). The duty cycle is sometimes referredto as D and is the percentage of a cycle of the current I_(L)(t) thatflows through the inductor L1. This variation in average current isprimarily due to variations of inductor ripple, which is referred to assub-harmonic oscillation, and slope compensation with duty cycle.

The converter 100 is susceptible to sub-harmonic oscillation for dutycycles greater than fifty percent in peak mode and duty cycles less thanfifty percent in valley mode. Peak mode is the configuration describedabove when the peak current in inductor L1 is limited and valley mode iswhen the valley current is limited. In order to attenuate thesub-harmonic oscillation, a compensating ramp, known as slopecompensation, is added to the inductor current sense signal I_(s)(t) inpeak current control mode or subtracted from the current sense signalI_(s)(t) in valley current mode control. The same current sense signalwith the added compensation ramp is typically presented to a currentlimit comparator because it is desirable to avoid sub-harmonicoscillation in either current limit mode.

When the converter 100 is based on buck topology, the output voltage Vagis less than the input voltage V_(IN). In the buck topology, the slopecompensation is achieved using a slope compensating ramp function. Theslope compensation is proportional to V_(OUT)/L in peak control and(V_(IN)-V_(OUT))/L in valley control. Depending on the input and outputconditions, the magnitude of the slope compensating ramp can dwarf theinductor current sense signal I_(S)(t), which often occurs in the caseof valley mode. The converters and methods described herein provide forcurrent limiting circuitry that is more accurate by removing the slopecompensation and avoiding sub-harmonic oscillation.

An embodiment of a DC to DC converter 200 that overcomes the problemsassociated with the prior art converter 100 is shown in FIG. 2. Theconverter 200 includes an input 202 that receives a DC input voltageV_(IN). The input voltage V_(IN) is converted to an output voltageV_(OUT) that is present on an output 204 of the converter 200. Theconverter 200 is configured as a boost converter, so the output voltageV_(OUT) is greater than the input voltage V_(IN). The converter 200includes an inductor L2 that is coupled between the input 202 and a nodeN1. The inductor L2 is sometimes referred to as having a first terminalthat is coupled to the input and a second terminal that is coupled tothe node N1. The inductor L2 stores energy and discharges energy in theform of electric current.

A switch SW1 is coupled between the node N1 and a current sensingresistor R1. The state of the switch SW1 is controlled by a pulse widthmodulation (PWM) controller, which is not shown in FIG. 1. The switchSW1 controls the charging and discharging of the inductor L2 along withthe duty cycle. A diode D2 is coupled between the node N1 and the output204. The diode D2 prevents the inductor L2 from charging by way ofvoltages present on the output 204. Therefore, the inductor L2 is onlyable to be charged from the input voltage V_(IN) at the input 202. Theoutput 204 is coupled to a low-pass filter 210, which in the embodimentof FIG. 2 includes a capacitor C2 and a resistor R2. The low-pass filter210 serves to attenuate ripple and transients.

The output voltage V_(OUT) is monitored by an error amplifier 212 thatamplifies the voltage difference between the output voltage V_(OUT) anda reference voltage V_(REF1). The output of the error amplifier 212 iscoupled to an input of a comparator 216. As described in greater detailbelow, the comparator 216 compares the output of the error amplifier 212to a voltage slope. The output of the comparator 216 is coupled to a PWMcontroller (not shown) that controls the state of the switch SW1 as isknown in the art.

The current sensing resistor R1 is connected to a node, which in theembodiment of FIG. 2 is ground. The resistor R1 is also coupled to afilter 218 that includes resistors R3 and R4 and a capacitor C3. Theresistance of the sensing resistor R1 is typically very small, so thevoltage generated by the sensing resistor R1 is amplified by anamplifier 220. The output of the amplifier 220 is coupled to an input ofa mixer 222. The mixer 222 has a second input that is coupled to a slopegenerator 224. The slope generator 224 generates a ramp function tooffset or attenuate the sub-harmonic oscillations when the duty cycleexceeds fifty percent for peak monitoring or is less than fifty percentfor valley monitoring. The output of the mixer 222 is coupled to thecomparator 216 where it is compared to the output of the error amplifier212 as described above.

The output of the slope generator 224 is also coupled to a sample andhold circuit 230 that includes a capacitor C4 and a switch SW2. Thesample and hold circuit 230 is coupled to the input of a mixer 240 whereit is subtracted from the output of the mixer 222. The waveformsassociated with the mixers 222 and 240 are described in greater detailbelow. The output of the mixer 240 is coupled to a comparator 242 whereit is compared to a predetermined voltage V_(REF2). The output of thecomparator 242 is a signal used to terminate the charging of theinductor L2. Accordingly, the voltage V_(REF2) is a voltage thatcorresponds to the maximum charging current that the inductor L2 can besubjected to. For example, if the inductor L2 is charging too high, itcan damage components in the converter 200 or generate an output voltageV_(OUT) that is too great.

The operation of the converter 200 will now be described. In summary, acombination of inductor current sense signal and slope compensation rampis presented to the current limit comparator 242. When this signalcrosses a predetermined threshold established by the voltage referenceV_(REF2), a current limit event is detected and a pulse used to closethe switch SW1 is immediately terminated for the rest of the cycle.Terminating the charging is accomplished by opening the switch SW1 sothat the inductor L2 cannot charge any further. The converter 200preserves the advantage of adding or deleting the slope compensationfrom the current sense signal using the mixer 222, which avoidssub-harmonic oscillation. The duty cycle that operates the switch SW1has a very small change from one cycle to the next cycle, so therequired slope compensation variation generated by the slope generator224 is also very small. Accordingly, the amount of slope compensationadded in a given cycle is peak detected and, using the sample and holdcircuit 230, the information is stored. At the beginning of thesubsequent cycle, the inductor current signal is offset by the slopecompensation information stored by the sample and hold circuit 230 fromthe previous cycle. Slope compensation is added or subtracted to thepresent cycle normally. Therefore, the current sense signal output bythe mixer 240 reaches the same peak as the previous cycle. This does notchange the onset of the inductor current limit signal output by thecomparator 242 and the presence of slope compensation will avoidsub-harmonic oscillation.

The operation of the converter 200 will now be described in greaterdetail. The converter 200 converts the input DC voltage V_(IN) to anoutput DC voltage V_(OUT). A PWM controller (not shown) closes theswitch SW1, which cause current to flow from the input 202, through theinductor L2, and through the sensing resistor R1. After a predeterminedtime, the switch SW1 is opened causing the energy stored in the inductorL1 to discharge current through the diode D2 and into the capacitor C2.Because the capacitor C2 and the resistor R2 form a low-pass filter, theoutput voltage V_(OUT) is a DC voltage.

The output voltage V_(OUT) is monitored by the error amplifier 212 whichamplifies the difference between the output voltage V_(OUT) and thereference voltage V_(REF1). The error amplifier 212 outputs a voltagethat is compared to the output of the mixer 222 by the comparator 216.if the voltage is low, the PWM controller adjusts the duty cycle of thepulses operating the switch SW1 to increase the charge on the inductorL2, which charges the capacitor C2 to a higher voltage. If the voltageis high, the PWM controller adjusts the duty cycle of the pulsesoperating the switch SW1 to decrease the charge on the inductor L2,which charges the capacitor C2 to a lower voltage. In other embodiments,the error amplifier 212 or other circuitry monitors the output voltageV_(OUT) to determine if it is within a predetermined range. The dutycycle of the pulses operating the switch SW1 is adjusted to bring theoutput voltage V_(OUT) within the predetermined specification.

The current used to charge the inductor L2 when the switch SW1 is closedis referred to as the inductor current and is sensed by the resistor R1.The value of the resistor R1 is very small so as not to interfere withthe charging of the inductor L2. The voltage generated by the inductorcurrent flowing through the resistor R1 is filtered by the filter 218and amplified by the amplifier 220. An example of the inductor currentis shown by the graph of FIG. 3A, which shows the time in which theinductor L2 is charging and discharging, wherein the charging portionhas a slope m1. The duty cycle is the ratio of the time that theinductor L2 is charging divided by the time of one period of a cycle, soit is related to the percentage of a cycle that the inductor L2 ischarging. The peak inductor current is referred to as X and is sometimesreferred to as the peak inductor current.

The inductor current is input to the mixer 222 where it is added to theslope compensation generated by the slope generator 224. An embodimentof the slope compensation generated by the slope generator 224 is shownin FIG. 3B. The slope compensation has a peak noted by Y, a durationthat corresponds to the charge time in FIG. 3A and a slope m2. The slopecompensation is also input to the sample and hold circuit 230 asdescribed in greater detail below. The mixer 222 outputs the sum of theinductor current and the slope compensation, which is shown by theexemplary graph of FIG. 3C. As shown in FIG. 3C, the peak output by themixer 222 is the sum of X and Y and the slope is also the sum of theslope m1 of the inductor current and the slope m2 of the slopecompensation. The output of the mixer 222 is input to the comparator 216where it is used to by the PWM controller (not shown) to control theswitch SW1 in a conventional manner.

The output of the mixer 222 is also output to the mixer 240 where thepeak of the slope compensation, FIG. 3B, is subtracted from the slopegenerator signal. The output of the mixer 240 is shown by the graph ofFIG. 3D. During the slope generation by the slope generator 224, theswitch SW2 is closed so that the capacitor C4 charges to the value of Y.After the peak has been reached, the switch SW2 is opened so that the DCvoltage of amplitude Y is input to the mixer 240. The output of themixer 240 has a peak X and a slope of m1 plus m2. The signal of FIG. 3Dis input to the comparator 242 where it is compared to the referencevoltage V_(REF2). If the voltage output by the mixer 240 exceeds thereference voltage V_(REF2), the comparator 242 outputs a signal thatterminates the charging of the inductor L2. More specifically, when thecomparator 242 outputs a voltage, the current limit of the inductor L2and/or the converter 200 has been reached, so the switch SW1 is opened.

The maximum slope compensation generated by the slope generator 224 ispeak detected from the previous cycle. This maximum slope compensationis offset or subtracted from the sum of the inductor current sensesignal and the slope compensation by the mixer 240. Therefore, the peakreached by the input to the comparator 242 is equal to X in the givencycle but the slope will have both the inductor current slope and theslope of the slope compensation in it. In conventional converters, thesignal to terminate the charging of the inductor is the sum of theinductor current and the slope compensation. In many situations, theslope compensation is a significant portion of this signal, so thecurrent limiting signal that terminates the charging of the inductor isnot accurate.

The advantages of the converter 200 over conventional converters aredescribed below. A combination of the inductor current signal and theslope compensation ramp generated by the slope generator 224 ispresented to the current limit comparator 242. When the signal crossesthe predetermined threshold set by the voltage reference V_(REF2), acurrent limit event is detected and the switch SW1 is opened in order toterminate the charging of the inductor L2 for rest of the cycle. Theconverter 200 preserves the advantage of adding or subtracting the slopecompensation from the inductor current signal, which preventssub-harmonic oscillation.

The duty cycle change from one cycle to the next is typically small andtherefore the required slope compensation variation from one cycle tothe next cycle is also small. Accordingly, the amount of slopecompensation added in a given cycle is peak detected and stored usingthe sample and hold circuit 230. At the beginning of the subsequentcycle, the inductor current signal is offset by the slope compensationinformation stored from the previous cycle in the sample and holdcircuit 230. The stored slope compensation is added or subtracted to thecurrent cycle by use of the mixer 240. Therefore, the signal input tothe comparator 242 signal will reach the same peak as the previouscycle. This will not modify the triggering of a signal from thecomparator 242 to limit inductor current; however, the presence of theslope compensation will avoid sub-harmonic oscillation.

The operation of the slope compensation with the inductor currentsensing is shown by the flowchart 400 of FIG. 4. The method of FIG. 4commences at step 402 with charging an inductor for a first period, theinductor being coupled between an input of the DC to DC converter andthe output of the DC to DC converter. In step 404, the inductor isdischarged for a second period, the combination of the first period andthe second period being a cycle in a frequency. The flowchart continuesat step 406 with generating an inductor current signal representative ofthe current flow through the inductor during a first period. In step 408a compensation signal is generated by adding a ramp function to theinductor current signal. In step 410 an offset signal is generated bysubtracting a value of the ramp function from the compensation signal.In step 412 the charging of the inductor is terminated in response tothe offset signal.

The converter 200 and the method of operating the converter 200 withregard to terminating the charging of the inductance L2 applies to buck,boost and buck-boost based DC to DC converters in peak current modecontrol, valley current mode control or voltage mode control with acurrent limit function.

Although illustrative embodiments have been shown and described by wayof example, a wide range of alternative embodiments is possible withinthe scope of the foregoing disclosure.

What is claimed is:
 1. A DC to DC converter comprising: an inductorhaving a first terminal and a second terminal, the first terminalcoupled to an input of the DC to DC converter and the second terminalbeing coupled to an output of the DC to DC converter; a first switch forcontrolling current flow through the inductor; a current sensor formeasuring the current flow through the inductor and generating aninductor current signal; a slope generator for generating a slopecompensation signal; a first mixer having an output, the first mixer foradding the slope compensation signal to the inductor current signal; asample and hold circuit for sampling a portion of the slope compensationsignal; and a second mixer having an output, the second mixer forsubtracting the sampled portion of the slope compensation signal fromthe output of the first mixer; wherein the first switch is operated inresponse to the output of the second mixer.
 2. The DC to DC converter ofclaim 1 further comprising a comparator for comparing the output of thesecond mixer to a predetermined value and wherein the first switch isoperated in response to the output of the second mixer.
 3. The DC to DCconverter of claim 1, wherein the sample and hold circuit is forsampling a peak value of the slope compensation signal.
 4. The DC to DCconverter of claim 1, wherein the first switch is coupled between theinductor and a ground.
 5. The DC to DC converter of claim 1, wherein thefirst switch has a closed state when current flows through the inductorand an open state when current does not flow through the inductor,wherein the first switch opens and closes per a frequency, wherein thesample and hold circuit samples a portion of the slope compensationoccurring during a first period of the frequency, and wherein the secondmixer mixes the sampled signal to the output of the first mixer based ona second period of the frequency.
 6. The DC to DC converter of claim 5wherein the second period of the frequency is the period directlyfollowing the first period.
 7. The DC to DC converter of claim 1,wherein the first switch is a field effect transistor.
 8. The DC to DCconverter of claim 1, wherein: the inductor current signal has a firstslope during a time when the inductor is charging; the slopecompensation signal has a second slope; and the output of the secondmixer has a third slope that is the sum of the first slope and thesecond slope.
 9. The DC to DC converter of claim 1, wherein the peakvoltage output by the second mixer has an amplitude that issubstantially the same as a peak amplitude of the inductor currentsignal.
 10. The DC to DC converter of claim 1, wherein the peak voltageoutput by the first mixer has an amplitude that is substantially the sumof the peak amplitude of the inductor current signal and the slopecompensation signal.
 11. The DC to DC converter of claim 1 furthercomprising: a second comparator for comparing the output voltage to apredetermined voltage; and a third comparator for comparing the outputof the second comparator to the output of the first mixer; wherein thestate of the first switch is at least partially controlled in responseto the output of the third comparator.
 12. The DC to DC converter ofclaim 1 further comprising a diode coupled between the inductor and theoutput.
 13. The DC to DC converter of claim 1 further comprising a lowpass filter coupled between the output and a ground potential.
 14. Amethod of operating a DC to DC converter, the method comprising:charging an inductor for a first period, the inductor being coupledbetween an input of the DC to DC converter and an output of the DC to DCconverter; discharging the inductor for a second period, the combinationof the first period and the second period being a cycle in a frequency;generating an inductor current signal representative of the current flowthrough the inductor during a first period; generating a compensationsignal by adding a slope compensation signal to the inductor currentsignal; generating an offset signal by subtracting a value of theamplitude of the slope compensation signal from the compensation signal;and terminating charging of the inductor in response to the offsetsignal.
 15. The method of claim 14, wherein the compensation signal hasa slope that is substantially equal to the sum of the inductor currentsignal and the slope compensation signal.
 16. The method of claim 14,wherein the compensation signal has a peak amplitude that issubstantially equal to the sum of the peak amplitude of the inductorcurrent signal and the peak amplitude of the slope compensation signal.17. The method of claim 14 further comprising storing the peak amplitudeof the slope compensation signal during a first cycle and whereingenerating an offset signal comprises subtracting a peak amplitude ofthe slope compensation signal from a subsequent cycle of thecompensation signal.
 18. The method of claim 14 wherein generating anoffset signal further comprises generating an offset signal that hassubstantially the same peak amplitude as the peak amplitude of theinductor current signal.
 19. The method of claim 14 further comprisingcomparing the offset signal to a predetermined voltage and whereinterminating charging of the inductor comprises terminating charging ofthe inductor in response to the offset signal being greater than thepredetermined voltage.
 20. A DC to DC converter comprising: an inductorhaving a first terminal and a second terminal, the first terminalcoupled to an input of the DC to DC converter and the second terminalbeing coupled to an output of the DC to DC converter; a first switchcoupled between the second terminal and a current sensor, the firstswitch having a closed state when current flows through the first switchand an open state when current does not flow through the first switch;the current sensor being coupled between the first switch and a ground,the current sensor for generating an inductor current signalrepresentative of the current flow through the sensor; a slope generatorfor generating a slope compensation signal; a sample and hold circuitfor storing the peak amplitude of the slope compensation signal during afirst period; a first mixer having an output, the first mixer for addingthe slope compensation signal to the inductor current signal; a secondmixer having an output, the second mixer for subtracting the sampledportion of the slope compensation signal from the output of the firstmixer during a second period, wherein the second period is subsequent tothe first period; and a comparator coupled to the output of the secondmixer, the comparator for comparing the output of the second mixer to apredetermined voltage; wherein the first switch is opened in response tothe output of the comparator.